High speed, low power comparator

ABSTRACT

A method for reducing bit errors in an analog to digital converter having an array of comparators. The outputs of first and second comparators are received as in inputs to an Exclusive OR gate. The first and second comparators are separated in the array by a third comparator. The output of the Exclusive OR gate is used to determine if the third comparator is in a metastable condition. If the third comparator is in a metastable condition, the bias current of the latch circuit of the third comparator is increased to increase the rate at which the third comparator transitions to a steady state.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No. 10/798,552, filed Mar. 12, 2004, now U.S. Pat. No. 6,876,318, which is a division of U.S. application Ser. No. 10/226,165, filed Aug. 23, 2002, now U.S. Pat. No. 6,727,839 issued Apr. 27, 2004.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to high speed, low power comparators.

2. Background Art

Commercialization of the Internet has proven to be a mainspring for incentives to improve network technologies. Development programs have pursued various approaches including strategies to leverage use of the existing Public Switched Telephone Network and plans to expand use of wireless technologies for networking applications. Both of these approaches (and others) entail the conversion of data between analog and digital formats. Therefore, it is expected that analog-to-digital converters (ADCs) and digital-to-analog converters (DACs) will continue to perform critical functions in many network applications.

Because ADCs find uses in a wide variety of applications, design of these circuits has evolved along many paths to yield several distinct architectures, including “delta sigma,” “successive approximation,” “pipelined,” “subranging,” “folding,” and “flash.” Comparators are the basic building block in each of these designs, and some architectures—such as pipelined, subranging, folding, and flash—use an array of comparators.

For example, FIG. 1 is a block diagram of an exemplary conventional two-bit flash ADC 100. ADC 100 comprises a first comparator “A” 102, a second comparator “B” 104, a third comparator “C” 106, a priority encoder 108, a first resistor “R₁” 110, a second resistor “R₂” 112, a third resistor “R₃” 114, and a fourth resistor “R₄” 116. Each of R₁ 110, R₂ 112, R₃ 114, and R₄ 116 has the same measure of resistance. R₁ 110, R₂ 112, R₃ 114, and R₄ 116 are connected in series between an analog ground “V_(AG)” 118 and a first supply voltage “V_(DD)” 120. (Alternatively, analog ground V_(AG) 118 can be replaced by a second supply voltage “V_(SS)”.) R₁ 110 is connected between V_(AG) 118 and a first node “N₁” 122. R₂ 112 is connected between N₁ 122 and a second node “N₂” 124. R₃ 114 is connected between N₂ 124 and a third node “N₃” 126. R₄ 116 is connected between N₃ 126 and V_(DD) 120. In this configuration, the voltage at N₁ 122 (the reference voltage of comparator A 102) is equal to V_(DD)/4, the voltage at N₂ 124 (the reference voltage of comparator B 104) is equal to V_(DD)/2, and the voltage at N₃ 126 (the reference voltage of comparator C 106) is equal to 3V_(DD)/4.

The inverting terminals of comparators A 102, B 104, and C 106 are connected to, respectively, N₁ 122, N₂ 124, and N₃ 126. An analog signal “x” 128 is received at an input 130, which is connected to the noninverting terminals of comparators A 102, B 104, and C 106. A quantized signal is produced at the output terminal of each comparator. Quantized signals “w₁” 132, “w₂” 134, and “w₃” 136 are produced at the output terminals of, respectively, comparators A 102, B 104, and C 106. Each quantized signal has a voltage with a value “LOW” or a value “HIGH” depending upon whether a corresponding value of the voltage of analog signal x 128 is less than (or equal to) or greater than the voltage at the inverting terminal of the corresponding comparator (i.e., the reference voltage of the comparator). For example, when the value of the voltage of analog signal x 128 is less than or equal to V_(DD)/4, the values of the voltages of w₃ 136, w₂ 134, and w₁ 132 are equal to, respectively, LOW, LOW, and LOW. When the value of the voltage of analog signal x 128 is less than or equal to V_(DD)/2, but greater than V_(DD)/4, the values of the voltages of w₃ 136, w₂ 134, and w₁ 132 are equal to, respectively, LOW, LOW, and HIGH. When the value of the voltage of analog signal x 128 is less than or equal to 3V_(DD)/4, but greater than V_(DD)/2, the values of the voltages of W₃ 136, w₂ 134, and w₁ 132 are equal to, respectively, LOW, HIGH, and HIGH. When the value of the voltage of analog signal x 128 is less than or equal to V_(DD), but greater than 3V_(DD)/4, the values of the voltages of w₃ 136, w₂ 134, and w₁ 132 are equal to, respectively, HIGH, HIGH, and HIGH.

The output terminals of comparators A 102, B 104, and C 106 are connected to priority encoder 108. Quantized signals w₁ 132, w₂ 134, and W₃ 136 are received by priority encoder 108, which processes them to produce, at an output 138, a two-bit digital signal “y” comprising a least significant bit (LSB) signal “y₁” 140 and a most significant bit (MSB) signal “y₂” 142.

The skilled artisan will appreciate that, with additional comparators and resistors and by using a priority encoder capable of processing additional quantized signals, flash ADC 100 can be modified so that digital signal y comprises more than two bit signals. Alternatively, flash ADC 100 can be modified so that digital signal y comprises one bit signal.

Implementations of comparators A 102, B 104, and C 106 often use current-mode latch circuits. FIG. 2 is a schematic diagram of an exemplary conventional current-mode latch circuit 200 that can be used in an implementation of any of comparators A 102, B 104, or C 106. Latch circuit 200 comprises a cross-connected pair of transistors 202 connected between a reset switch 204 and first supply voltage V_(SS) 118. Preferably, cross-connected pair 202 comprises a first NMOSFET (n-channel Metal Oxide Semiconductor Field Effect Transistor) “M₁” 206 and a second NMOSFET “M₂” 208. Ideally, M₁ 206 and M₂ 208 are matched transistors. Preferably, each of M₁ 206 and M₂ 208 has a gain greater than one. However, cross-connected pair 202 can function if the product of the individual gains of M₁ 206 and M₂ 208 (i.e., the loop gain) is greater than one. The gate terminal of M₂ 208 is connected to the drain terminal of M₁ 206 at a first port “N₄” 210. The gate terminal of M₁ 206 is connected to the drain terminal of M₂ 208 at a second port “N₅” 212. The source terminals of M₁ 206 and M₂ 208 are together connected to analog ground V_(AG) 118. Preferably, reset switch 204 comprises a third NMOSFET “M₃” 214. The source terminal of M₃ 214 is connected to the drain terminal of one of M₁ 206 or M₂ 208; the drain terminal of M₃ 214 is connected the drain terminal of the other of M₁ 206 or M₂ 208. A clock waveform “Ck” 216 is applied to the gate terminal of M₃ 214. Ck 216 cycles between an “UP” voltage and an “DOWN” voltage at a sampling frequency.

The skilled artisan will appreciate that M₁ 206, M₂ 208, and M₃ 214 can also be realized in other field effect, junction, or combination transistor technologies. In general, reset switch 204 can be realized in a variety of switch technologies, including microelectromechanical embodiments. Latch circuit 200 can also be used for other applications.

For each latch circuit 200 in ADC 100, quantized signal “w” (e.g., w₁ 132, w₂ 134, or w₃ 136) is produced as an output voltage at N₄ 210 or N₅ 212. Latch circuit 200 is often preceded by an input stage (not shown) that includes a differential amplifier so that the voltage of analog signal x 128, applied at the noninverting terminal of the comparator, can be compared with the voltage at the inverting terminal of the comparator. For example, the voltage of analog signal x 128 is compared with V_(DD)/4, for comparator A 102; V_(DD)/2, for comparator B 104; and 3V_(DD)/4, for comparator C 106.

For each latch circuit 200 in ADC 100, the input stage produces a differential current signal comprising a first current signal “i₁” 218 and a second current signal “i₂” 220. First and second current signals i₁ 218 and i₂ 220 each comprise a bias current “i_(b)” and a signal current “i_(s)”. The relationship between bias current i_(b) and signal current i_(s) in first current signal i₁ 218 can be expressed as shown in Eq. (1): i ₁ =i _(b)+(½)(i _(s)),  Eq. (1) while the relationship between bias current i_(b) and signal current i_(s) in second current signal i₂ 220 can be expressed as shown in Eq. (2): i ₂ =i _(b)−(½)(i _(s)).  Eq. (2) The differential amplifier is configured so that first current signal i₁ 218 increases and decreases in response to, respectively, the rise and drop of the voltage of analog signal x 128, while second current signal i₂ 220 increases and decreases in response to, respectively, the drop and rise of the voltage of analog signal x 128. Thus, first and second current signals i₁ 218 and i₂ 220 always change currents in opposite directions, but the sum of first and second current signals i₁ 218 and i₂ 220 remains equal to twice the bias current i_(b).

For each latch circuit 200 in ADC 100, the differential amplifier is configured so that no signal current i_(s) is produced when the voltage of analog signal x 128, applied at the noninverting terminal of the comparator, equals the voltage at the inverting terminal of the comparator. For example, for comparator A 102, no signal current i_(s) is produced when the voltage of analog signal x 128 equals V_(DD)/4; for comparator B 104, no signal current i_(s) is produced when the voltage of analog signal x 128 equals V_(DD)/2; and for comparator C 106, no signal current i_(s) is produced when the voltage of analog signal x 128 equals 3V_(DD)/4.

In latch circuit 200, first current signal i₁ 218 and second current signal i₂ 220 are received as input current signals at, respectively, N₄ 210 and N₅ 212. When the voltage of Ck 216 is UP (i.e, the reset phase), M₃ 214 connects N₄ 210 with N₅ 212, so that the steady state voltages at both nodes are equal, and bias current i_(b) flows through each of M₁ 206 and M₂ 208. Parasitic capacitances at each of nodes N₄ 210 and N₅ 212 are charged by bias current i_(b) that flows through each of M₁ 206 and M₂ 208. The skilled artisan will appreciate that the parasitic capacitance at, for example, N₄ 210, includes the gate-to-source capacitance of M₂ 208, the drain-to-substrate capacitance of M₁ 206, the drain-to-substrate capacitance of M₃ 214, and the capacitance of the wiring connecting circuit devices. Bias current i_(b) charges the parasitic capacitances at each of nodes N₄ 210 and N₅ 212 so that the voltages at N₄ 210 and N₅ 212 are at a metastable “MID” value that is between LOW and HIGH. The gate and drain terminals of M₁ 206 and M₂ 208 are connected together. M₁ 206 and M₂ 208 are sized so that, under these conditions, they operate in “ON” states.

When the voltage of Ck 216 is DOWN (i.e., the sampling phase), the states of M₁ 206 and M₂ 208 are controlled by first and second current signals i₁ 218 and i₂ 220. For example, when first current signal i₁ 218 is greater than bias current i_(b) and second current signal i₂ 220 is less than bias current i_(b), a transient is initiated to force M₁ 206 to operate in an “OFF” state, while M₂ 208 remains operating in an ON state. The course of this transient depends on how first and second current signals i₁ 218 and i₂ 220 change during the sampling phase. If M₁ 206 is turned OFF and the parasitic capacitances at N₄ 210 are fully charged by first current signal i₁ 218 (i.e., at a new steady state), the voltage at N₄ 210 is HIGH and the voltage at N₅ 212 is LOW.

It is a characteristic of latch circuit 200 that the port (i.e., N₄ 210 or N₅ 212) receiving the current signal (i.e., i₁ 218 or i₂ 220) that is greater than bias current i_(b) requires more time to reach its new steady state voltage than the port receiving the current signal that is less than bias current i_(b). However, if first and second current signals i₁ 218 and i₂ 220 both have values near to that of bias current i_(b) (i.e., small signal current i_(s)), it is possible that the output voltage (at N₄ 210 or N₅ 212) may not reach LOW or HIGH before the end of the sampling phase, but remain in a metastable condition. Such a situation is more likely to occur if Ck 216 cycles at a high sampling frequency. In this situation, the quantized signal (i.e., w₁ 132, w₂ 134, or w₃ 136) produced by the comparator associated with latch circuit 200 (i.e., comparator A 102, B 104, or C 106) does not registered as a digital input to priority encoder 108. Consequently, ADC 100 does not produce a digital signal y. Such a “non-decision” is referred to as a “bit error”. Bit errors can adversely effect the performance of a system that uses the digital output of ADC 100.

Bit errors can be reduced by increasing bias current i_(b) so that only a small signal current i_(s) is needed to force the port (i.e., N₄ 210 or N₅ 212) receiving the current signal (i.e., i₁ 218 or i₂ 220) that is greater than bias current i_(b) to reach its new steady state voltage. This increases the overall speed of latch circuit 200. However, increasing bias current i_(b) can decrease the signal-to-noise ratio of ADC 100. Moreover, increasing bias current i_(b) in all of the comparators of ADC 100 causes ADC 100 to dissipate more power, particularly because each comparator draws twice the bias current i_(b) during both the sampling and the reset phases. Such a situation is undesirable where ADC 100 is employed in a system that demands low power consumption, such as a portable wireless application. What is needed is a technique to identify which comparator, in the array of comparators, is in a metastable condition, and to increase the rate at which the identified comparator transitions to a steady state.

BRIEF SUMMARY OF THE INVENTION

The present invention relates to high speed, low power comparators. In an array of comparators, the present invention provides a technique to identify which comparator is in a metastable condition, and to increase the rate at which the identified comparator transitions to a steady state. A bias current is provided to the identified comparator in the metastable condition, such that the rate at which the comparator in the metastable condition transitions to the steady state is increased.

In an embodiment, the bias current is provided by controlling a current output from a variable current source that provides the bias current for a latch circuit of the identified comparator in the metastable condition.

In another embodiment, the comparator in the metastable condition is identified by comparing a characteristic of a first comparator of the array of comparators with a characteristic of a second comparator of the array of comparators. The first comparator and the second comparator are separated in the array of comparators by a third comparator in the array of comparators. It is determined if the third comparator is the comparator in the metastable condition based on the compared characteristics. Preferably, the characteristics are compared by receiving the first and second characteristics as inputs to an Exclusive OR gate.

In yet another embodiment, the bias current is provided by controlling a current output from a variable current source that provides the bias current for a latch circuit of the identified comparator in the metastable condition with an output of an Exclusive OR gate.

In still another embodiment, the bias current is provided by connecting a first current source in parallel with a second current source to increase the bias current for a latch circuit of the identified comparator in the metastable condition. Preferably, a switch that connects the first current source in parallel with the second current source is controlled by an output of an Exclusive OR gate.

The present invention also provides a method to increase, in an array of comparators that includes a first, a second, and a third comparator, a rate at which the third comparator transitions to a steady state. An output of the first comparator is compared with an output of the second comparator, and a bias current is provided to the third comparator based on the compared first and second outputs.

In an embodiment, the outputs are compared by receiving the first and second outputs as inputs to an Exclusive OR gate. Preferably, a variable current source that provides the bias current for a latch circuit of the third comparator is controlled based on an output of an Exclusive OR gate.

In another embodiment, the bias current is provided to the third comparator by connecting a first current source in parallel with a second current source to increase the bias current for a latch circuit of the third comparator. Preferably, a switch that connects the first current source in parallel with the second current source is controlled based on an output of an Exclusive OR gate.

The present invention also comprises an array of comparators comprising a first, a second, and a third comparator, an Exclusive OR gate having a first input connected to an output of the first comparator and a second input connected to an output of the second comparator, and a variable current source connected to an output of the Exclusive OR gate. The variable current source supplies a bias current to the third comparator. Preferably, the output of the Exclusive OR gate produces a signal that controls the variable current source. Preferably, the third comparator is arranged in the array of comparators between the first comparator and the second comparator.

In an embodiment, the third compararator comprises a latch circuit configured to receive the bias current. Preferably, the latch circuit comprises a cross connected pair of transistors connected between a reset switch and a supply voltage. The latch circuit has a first port capable of receiving a first current signal and producing a first output voltage, and a second port capable of receiving a second current signal and producing a second output voltage. In an embodiment, the cross connected pair of transistors comprises a first MOSFET and a second MOSFET configured so that the gate terminal of the first MOSFET is connected to the drain terminal of the second MOSFET, the gate terminal of the second MOSFET is connected to the drain terminal of said the MOSFET, and the source terminals of the first and the second MOSFETs are connected to the supply voltage. Preferably, the reset switch comprises a MOSFET connected between the first port and the second port.

In another embodiment, the array of comparators further comprises a second Exclusive OR gate having an input connected to an output of the third comparator, and a second variable current source connected to an output of the second Exclusive OR gate. The second variable current source supplies a second bias current to the second comparator.

The present invention also comprises an analog to digital converter. The analog to digital comparator comprises an array of comparators, a priority encoder, an array of Exclusive OR gates, and an array of variable current sources. The array of comparators has respective inputs configured to receive an analog signal, and respective outputs configured to produce quantized signals responsive to the analog signal. The priority encoder is connected to the array of comparators, and is configured to produce a digital signal at an output responsive to the quantized signals. Each Exclusive OR gate of the array of Exclusive OR gates is configured to receive two of the quantized signals. Each variable current source of the array of variable current sources is configured to provide a bias current to a corresponding comparator of the array of comparators, and is controlled by an output of a corresponding Exclusive OR gate of the array of Exclusive OR gates.

In an embodiment, each Exclusive OR gate of the array of Exclusive OR gates produces a logic signal that controls a corresponding variable current source of the array of variable current sources. Preferably, each comparator of the array of comparators includes a latch circuit configured to receive a corresponding bias current. In another embodiment, the corresponding bias current is capable of being increased by a corresponding variable current source of the array of variable current sources.

Further embodiments, features, and advantages of the present invention, as well as the structure and operation of the various embodiments of the present invention, are described in detail below with reference to the accompanying figures.

BRIEF DESCRIPTION OF THE FIGURES

The accompanying drawings, which are incorporated herein and form part of the specification, illustrate the present invention and, together with the description, further serve to explain the principles of the invention and to enable a person skilled in the pertinent art to make and use the invention.

FIG. 1 is a block diagram of an exemplary conventional two-bit flash ADC 100.

FIG. 2 is a schematic diagram of an exemplary conventional current-mode latch circuit 200 that can be used in an implementation of any of comparators A 102, B 104, or C 106.

FIG. 3 is a block diagram of an array 300 of current-mode comparators configured in the manner of an embodiment of the present invention.

FIG. 4A is a schematic diagram of an input stage 400 that can be used with latch circuit 200 in an implementation of any of comparators A 102, B 104, or C 106.

FIG. 4B is a schematic diagram of an input stage 450 that can be used with latch circuit 200 in an implementation of any of comparators A 102, B 104, or C 106.

FIG. 5 is a block diagram of a portion of an array 500 of current-mode comparators configured in the manner of another embodiment of the present invention.

FIG. 6 is a block diagram of a portion of an array 600 of current-mode comparators configured in the manner of yet another embodiment of the present invention.

FIG. 7 is a schematic diagram of another current-mode latch circuit 700 that can be used in a realization of a comparator of the present invention.

FIG. 8 shows a flow chart of a method 800 for increasing, in an array of comparators, a rate at which a comparator in a metastable condition transitions to a steady state.

FIG. 9 shows a flow chart of a preferred method to identify the comparator in the metastable condition.

FIG. 10 shows a flow chart of a method 1000 for increasing, in an array of comparators that includes a first, a second, and a third comparator, a rate at which the third comparator transitions to a steady state.

The preferred embodiments of the invention are described with reference to the figures where like reference numbers indicate identical or functionally similar elements. Also in the figures, the left-most digit of each reference number identifies the figure in which the reference number is first used.

DETAILED DESCRIPTION OF THE INVENTION

The present invention relates to high speed, low power comparators. Where a functional component of a system—such as, but not limited to, a pipelined, subranging, folding, or flash ADC—uses an array of comparators, the present invention provides a technique to identify which comparator is in a metastable condition, and to increase the rate at which the identified comparator transitions to a steady state.

FIG. 3 is a block diagram of an array 300 of current-mode comparators configured in the manner of an embodiment of the present invention. Array 300 comprises comparator A 102, comparator B 104, comparator C 106, an Exclusive OR gate “XOR” 302, and a variable current source “I_(v)” 304. Quantized signals w₁ 132 and w₃ 136 are received as inputs to XOR 302. XOR 302 produces a logic signal “s” 306 that controls variable current source I_(v) 304. Variable current source I_(v) 304 augments bias current i_(b) for the latch circuit associated with second comparator B 104 in response to the value of logic signal s 306.

The skilled artisan will appreciate that logic signal s 306 equals one only if quantized signals w₁ 132 and w₃ 136 have different values. If quantized signals w₁ 132 and w₃ 136 have the same values, then logic signal s 306 equals zero. For example, when the values of the voltages of w₁ 132 and w₃ 136 are equal to, respectively, LOW and LOW, then logic signal s 306 is zero. When the values of the voltages of w₁ 132 and w₃ 136 are equal to, respectively, LOW and HIGH, then logic signal s 306 is one. When the values of the voltages of w₁ 132 and w₃ 136 are equal to, respectively, HIGH and LOW, then logic signal s 306 is one. When the values of the voltages of w₁ 132 and w₃ 136 are equal to, respectively, HIGH and HIGH, then logic signal s 306 is zero. The skilled artisan will also appreciate that such a comparison of quantized signals w₁ 132 and w₃ 136 can be realized by applying them to other types of logic gates that are configured in a manner to produce the same result as XOR 302.

The present invention is based on the likelihood that, for example, comparator B 104 will be in a metastable condition when comparator A 102 produces quantized signal w₁ 132 with value HIGH and comparator C 106 produces quantized signal w₃ 136 with value LOW. In this situation, logic signal s 306 is one and, in response, variable current source I_(v) 304 augments bias current i_(b) for the latch circuit associated with comparator B 104. Increasing bias current i_(b) increases both first and second current signals i₁ 218 and i₂ 220 and decreases the time needed for the port (i.e., N₄ 210 or N₅ 212) receiving the current signal (i.e., i₁ 218 or i₂ 220) that is greater than bias current i_(b) to reach its new steady state voltage. This decreases the probability that comparator B 104 will remain in a metastable condition and thus reduces the bit error rate (BER).

For example, in ADC 100, if analog signal x 128 is nearly equal to V_(DD)/2, then a small signal current i_(s) is produced for comparator B 104, a large positive signal current i_(s) is produced for comparator A 102, and a large negative signal current i_(s) is produced for comparator C 106. In this situation, comparator A 102 quickly produces quantized signal w₁ 132 with value HIGH, and comparator C 106 quickly produces quantized signal w₃ 136 with value LOW, but comparator B 104 may be slow to produce a digital value for quantized signal w₂ 134 before the end of the sampling phase. Increasing bias current i_(b) to the latch circuit associated with comparator B 104 increases its overall speed, decreases the likelihood that it will remain in a metastable state, and reduces the BER.

FIG. 4A is a schematic diagram of an input stage 400 that can be used with latch circuit 200 in an implementation of any of comparators A 102, B 104, or C 106. Input stage 400 receives analog signal x 128 and produces first and second current signals i₁ 218 and i₂ 220, which are received by latch circuit 200. Input stage 400 comprises amplifying MOSFETs “M₄” 402 and “M₅” 404, load MOSFETs “M₆” 406 and “M₇” 408, current mirror MOSFETs “M₈” 410 and “M₉” 412, and variable current source I_(v) 304.

Amplifying MOSFETs M₄ 402 and M₅ 404 are configured as a differential pair with their source terminals connected together. A load MOSFET is connected to the drain terminal of each amplifying MOSFET. The drain terminal of M₆ 406 is connected to the drain terminal of M₄ 402; the drain terminal of M₇ 408 is connected to the drain terminal of M₅ 404. The source terminals of M₆ 406 and M₇ 408 are together connected to first supply voltage V_(DD) 120. The source terminals of current mirror MOSFETs M₈ 410 and M₉ 412 are also together connected to first supply voltage V_(DD) 120. The gate terminal of M₈ 410 is connected to the gate and drain terminals of M₆ 406; the gate terminal of M₉ 412 is connected to the gate and drain terminals of M₇ 408. Variable current source I_(v) 304 is connected between the source terminal of M₄ 402 and M₅ 404 and analog ground V_(AG) 118. In input stage 400, M₄ 402 and M₅ 404 are NMOSFETs, while M₆ 406, M₇ 408, M₉ 410, and M₉ 412 are PMOSFETs (p-channel MOSFETs). However, this configuration can be reversed depending upon the overall configuration of the comparator associated with latch circuit 200. Furthermore, the skilled artisan will appreciate that M₄ 402, M₅ 404, M₆ 406, M₇ 408, M₈ 410, and M₉ 412 can also be realized in other field effect, junction, or combination transistor technologies.

The voltage of analog signal x 128 is received by input stage 400 at the noninverting terminal of the comparator (e.g., A 102, B 104, or C 106). This allows the voltage of analog signal x 128 to be compared with a reference voltage “ref” 414 received at the inverting terminal of the comparator. For example, the voltage of analog signal x 128 is compared with V_(DD)/4, for comparator A 102; V_(DD)/2, for comparator B 104; and 3 V_(DD)/4, for comparator C 106. The noninverting terminal of the comparator is connected to the gate terminal of M₄ 402. The inverting terminal of the comparator is connected to the gate terminal of M₅ 404.

Amplifying MOSFETs M₄ 402 and M₅ 404 act to control the distribution of current provided by variable current source I_(v) 304. The sum of the current flowing through both M₄ 402 and M₅ 404 equals the current provided by variable current source I_(v) 304, which is equal to twice bias current i_(b). For example, as the voltage received at the gate terminal of M₄ 402 rises with respect to the voltage received at the gate terminal of M₅ 404, the portion of the total current that flows through M₄ 402 and M₆ 406 increases, while the portion of the total current that flows through M₅ 404 and M₇ 408 decreases. M₈ 410 mirrors the increase in current flowing through M₆ 406 to produce first current signal i₁ 218 at the drain terminal of M₈ 410. M₉ 412 mirrors the decrease in current flowing through M₇ 408 to produce second current signal i₂ 120 at the drain terminal of M₉ 412.

Variable current source I_(v) 304 is controlled by Exclusive OR gate XOR 302. When logic signal s 306 produced by Exclusive OR gate XOR 302 is one, the current produced by variable current source I_(v) 304 is increased, which increases bias current i_(b) for latch circuit 200. Increasing bias current i_(b) increases both first and second current signals i₁ 218 and i₂ 220 and decreases the time needed for the port (i.e., N₄ 210 or N₅ 212) receiving the current signal (i.e., i₁ 218 or i₂ 220) that is greater than bias current i_(b) to reach its new steady state voltage. This increases the overall speed of latch circuit 200 and decreases the likelihood that it will remain in a metastable state.

FIG. 4B is a schematic diagram of an input stage 450 that can be used with latch circuit 200 in an implementation of any of comparators A 102, B 104, or C 106. Input stage 450 is configured in the same manner as input stage 400 except that: (1) a fixed current source “2 i _(b)” 416 is connected in parallel with variable current source I_(v) 304 between the source terminal of M₄ 402 and M₅ 404 and analog ground V_(AG) 118, and (2) a switch “S” 418 is connected in series with variable current source I_(v) 304 between the source terminal of M₄ 402 and M₅ 404 and analog ground V_(AG) 118.

Fixed current source 2 i _(b) 416 produces a current that is equal to twice bias current i_(b). Switch S 418 is controlled by Exclusive OR gate XOR 302. When logic signal s 306 produced by Exclusive OR gate XOR 302 is zero, switch S 418 is opened; when logic signal s 306 produced by Exclusive OR gate 302 is one, switch S 418 is closed. When switch S 418 is closed, the sum of the current flowing through both M₄ 402 and M₅ 404 equals the current provided by the sum of fixed current source 2 i _(b) 416 and variable current source I_(v) 304. This sum current increases both first and second current signals i₁ 218 and i₂ 220 and decreases the time needed for the port (i.e., N₄ 210 or N₅ 212) receiving the current signal (i.e., i₁ 218 or i₂ 220) that is greater than bias current i_(b) to reach its new steady state voltage. This increases the overall speed of latch circuit 200 and decreases the likelihood that it will remain in a metastable state.

As input stages 400 and 450 demonstrate, the skilled artisan could conceive of any number of circuits that could increase bias current i_(b) based on the teachings given herein. Therefore, the present invention is not limited to the teachings of input stages 400 and 450.

FIG. 5 is a block diagram of a portion of an array 500 of current-mode comparators configured in the manner of another embodiment of the present invention. The portion of array 500 comprises a comparator “O” 502, comparator A 102, comparator B 104, comparator C 106, a comparator “D” 504, a comparator “E” 506, an Exclusive OR gate “XOR_(A)” 508, Exclusive OR gate XOR 302, an Exclusive OR gate “XOR_(C)” 510, an Exclusive OR gate “XOR_(D)” 512, a variable current source “I_(vO)” 514, a variable current source “I_(vA)” 516, variable current source I_(v) 304, a variable current source “I_(vC)” 518, a variable current source “I_(vD)” 520, and a variable current source “I_(vE)” 522.

As with array 300, the comparators, Exclusive OR gates, and variable current sources of portion of array 500 are configured such that an Exclusive OR gate produces a logic signal that controls a variable current source that augments bias current for a latch circuit of a comparator of the array. The Exclusive OR gate receives as inputs quantized signals from other comparators of the array that are adjacent on either side of the bias current augmented comparator. Thus, the portion of array 500 expands upon the teachings of array 300 to show how the present invention operates in an environment of multiple Exclusive OR gates.

For example, if analog signal x 128 is nearly equal to the reference voltage of comparator B 104, then comparators O 502 and A 102 will quickly produce quantized signals with values HIGH, and comparators C 106, D 504, and E 506 will quickly produce quantized signals with values LOW, but comparator B 104 may be slow to produce a digital value for its quantized signal before the end of the sampling phase.

In this situation, XOR_(A) 508, which receives inputs from comparators O 502 and B 104, does not produce a digital output; XOR 302, which receives inputs from comparators A 102 and C 106, produces a digital output of one; XOR_(C) 510, which receives its input from comparators B 104 and D 504, does not produce a digital output; and XOR_(D) 512, which receives inputs from comparators C 106 and E 506, produces a digital output of zero. Thus, XOR 302 acts to cause variable current source I_(v) 304 to augment bias current i_(b) for the latch circuit associated with comparator B 104.

If, in response to an increase in bias current i_(b) for the latch circuit associated with comparator B 104, comparator B 104 transitions to a new steady state of, for example, HIGH, then XOR_(A) 508 produces a digital output of zero, and XOR_(C) 510 produces a digital output of one. Thus, XOR_(C) 510 acts to cause variable current source I_(VC) 518 to augment bias current i_(b) for the latch circuit associated with comparator C 106.

The remaining Exclusive OR gates do not cause their respective variable current sources to augment the bias currents for the latch circuits associated with their comparators. Advantageously, this: (1) increases the speed of comparator B 104 and decreases the likelihood that it will remain in a metastable state, (2) reduces the BER of an ADC realized with array 500, and (3) limits the increase in current drawn (and hence power dissipated) by array 500 to realize the increased speed of comparator B 104.

FIG. 6 is a block diagram of a portion of an array 600 of current-mode comparators configured in the manner of yet another embodiment of the present invention. The portion of array 600 comprises comparator O 502, comparator A 102, comparator B 104, comparator C 106, comparator D 504, comparator E 506, an Exclusive OR gate “XOR_(AB)” 602, an Exclusive OR gate “XOR_(BC)” 604, an Exclusive OR gate “XOR_(CD)” 606, variable current source I_(vO) 514, variable current source I_(vA) 516, variable current source I_(v) 304, variable current source I_(vC) 518, variable current source I_(vD) 520, and variable current source I_(vE) 522.

The portion of array 600 expands upon the teachings of array 500 to show how the present invention can provide the designer with a tradeoff between power dissipated and die area consumed. In the portion of array 600, XOR_(AB) 602 receives inputs from comparators O 502 and C 106, and controls variable current sources I_(vA) 516 and I_(v) 304; XOR_(BC) 604 receives inputs from comparators A 102 and D 504, and controls variable current sources I_(v) 304 and I_(vC) 518; and XOR_(CD) 606 receives inputs from comparators B 104 and E 506, and controls variable current sources I_(vC) 518 and I_(vD) 520.

In this configuration, for example, if analog signal x 128 is nearly equal to the reference voltage of comparator B 104, then comparators O 502 and A 102 will quickly produce quantized signals with values HIGH, and comparators C 106, D 504, and E 506 will quickly produce quantized signals with values LOW, but comparator B 104 may be slow to produce a digital value for its quantized signal before the end of the sampling phase.

In this situation, XOR_(AB) 602 produces a digital output of one, XOR_(BC) 604 produces a digital output of one, and XOR_(CD) 606 does not produce a digital output. Thus, XOR_(AB) 602 and XOR_(BC) 604 act to cause variable current sources I_(vA) 516, I_(v) 304, and IvC 518 to augment bias currents i_(b) for the latch circuits associated with comparators A 102, B 104, and C 106.

If, in response to an increase in bias current i_(b) for the latch circuit associated with comparator B 104, comparator B 104 transitions to a new steady state of, for example, HIGH, then XOR_(CD) 606 produces a digital output of one. Thus, XOR_(CD) 606 acts to cause variable current source I_(VD) 520 to augments bias current i_(b) for the latch circuit associated with comparator D 504.

The remaining Exclusive OR gates do not cause their respective variable current sources to augment the bias currents for the latch circuits associated with their comparators. Thus, for comparable realizations of arrays 500 and 600, array 600 draws more current (and hence dissipates more power) than array 500. However, because array 600 uses fewer Exclusive OR gates, array 600 consumes less die area than array 500.

FIG. 7 is a schematic diagram of another current-mode latch circuit 700 that can be used in a realization a comparator of the present invention. Latch circuit 700 comprises latch circuit 200, a first vertical latch 702 with a first vertical latch reset switch 704, a second vertical latch 706 with a second vertical latch reset switch 708, and a second pair of cross connected transistors 710.

First vertical latch 702 is connected between analog ground V_(AG) 118 and first supply voltage V_(DD) 120. Preferably, first vertical latch 702 comprises a fourth NMOSFET “M₁₀” 712 and a first PMOSFET “M₁₁” 714, where M₁₀ 712 and M₁₁ 714 are matched transistors. Preferably, each of M₁₀ 712 and M₁₁ 714 has a gain greater than one. However, first vertical latch 702 can function if the product of the individual gains of M₁₀ 712 and M₁₁ 714 (i.e., the loop gain) is greater than one. The source terminal of M₁₀ 712 is connected to analog ground V_(AG) 118. The drain terminal of M₁₀ 712 is connected to the gate terminal of M₁₁ 714. The gate terminal of M₁₀ 712 is connected to the gate terminal of M₂ 208. The source terminal of M₁₁ 714 is connected to first supply voltage V_(DD) 120. The drain terminal of M₁₁ 714 is connected to the gate terminal of M₁₀ 712. The skilled artisan will appreciate that M₁₀ 712 and M₁₁ 714 can also be realized in other field effect, junction, or combination transistor technologies.

Preferably, first vertical latch reset switch 704 comprises a second PMOSFET “M₁₂” 716. The source terminal of M₁₂ 716 is connected to first supply voltage V_(DD) 120. The drain terminal of M₁₂ 716 is connected to the gate terminal of M₁₁ 714. An inverse clock waveform “Ck.bar” 718 is applied to the gate terminal of M₁₂ 716. Ck.bar 718 cycles between DOWN voltage and UP voltage at the sampling frequency in a manner such that when the voltage of Ck 216 is UP, the voltage of Ck.bar 718 is DOWN, and vice versa. The skilled artisan will appreciate that M₁₂ 716 can also be realized in other field effect, junction, or combination transistor technologies. In general, first vertical latch reset switch 704 can be realized in a variety of switch technologies, including microelectromechanical embodiments.

Second vertical latch 706 is connected between analog ground V_(AG) 118 and first supply voltage V_(DD) 120. Preferably, second vertical latch 706 comprises a fifth NMOSFET “M₁₃” 720 and a third PMOSFET “M₁₄” 722, where M₁₃ 720 and M₁₄ 722 are matched transistors. Preferably, each of M₁₃ 720 and M₁₄ 722 has a gain greater than one. However, second vertical latch 706 can function if the product of the individual gains of M₁₃ 720 and M₁₄ 722 (i.e., the loop gain) is greater than one. The source terminal of M₁₃ 720 is connected to analog ground V_(AG) 118. The drain terminal of M₁₃ 720 is connected to the gate terminal of M₁₄ 722. The gate terminal of M₁₃ 720 is connected to the gate terminal of M₁ 206. The source terminal of M₁₄ 722 is connected to first supply voltage V_(DD) 120. The drain terminal of M₁₄ 722 is connected to the gate terminal of M₁₃ 720. The skilled artisan will appreciate that M₁₃ 720 and M₁₄ 722 can also be realized in other field effect, junction, or combination transistor technologies.

Preferably, second vertical latch reset switch 708 comprises a fourth PMOSFET “M₁₅” 724. The source terminal of M₁₅ 724 is connected to first supply voltage V_(DD) 120. The drain terminal of M₁₅ 724 is connected to the gate terminal of M₁₄ 722. Inverse clock waveform Ck.bar 506 is applied to the gate terminal of M₁₅ 724. The skilled artisan will appreciate that M₁₃ 720, M₁₄ 722, and M₁₅ 724 can also be realized in other field effect, junction, or combination transistor technologies. In general, second vertical latch reset switch 708 can be realized in a variety of switch technologies, including microelectromechanical embodiments.

Preferably, second cross connected pair 710 comprises a fifth PMOSFET “M₁₆” 726 and a sixth PMOSFET “M₁₇” 728, where M₁₆ 726 and M₁₇ 728 are matched transistors. Preferably, each of M₁₆ 726 and M₁₇ 728 has a gain greater than one. However, second cross connected pair 710 can function if the product of the individual gains of M₁₆ 726 and M₁₇ 728 (i.e., the loop gain) is greater than one. The gate terminal of M₁₇ 728 is connected to the drain terminal of M₁₆ 726 and to the gate terminal of M₁₄ 722. The gate terminal of M₁₆ 726 is connected to the drain terminal of M₁₇ 728 and to the gate terminal of M₁₁ 714. The source terminals of M₁₆ 726 and M₁₇ 728 are together connected to first supply voltage V_(DD) 120. The skilled artisan will appreciate that M₁₆ 726 and M₁₇ 728 can also be realized in other field effect, junction, or combination transistor technologies.

First vertical latch 702 and second vertical latch 706 act to increase the rate at which the port (i.e., N₄ 210 or N₅ 212) receiving the current signal (i.e., i₁ 218 or i₂ 220) that is greater than bias current i_(b) reaches its new steady state voltage.

For example, when the voltage of Ck 216 is DOWN (i.e., the sampling phase), the states of M₁ 206 and M₂ 208 are controlled by first and second current signals i₁ 218 and i₂ 220. If first current signal i₁ 218 is greater than bias current i_(b), first current signal i₁ 218 continues to charge the parasitic capacitances at N₄ 210, which causes the voltage at N₄ 210 to rise. Contemporaneously, when second current signal i₂ 220 is less than bias current i_(b), the parasitic capacitances at N₅ 212 start to discharge, which causes the voltage at N₅ 212 to drop.

Because the voltage at N₅ 212 is also the voltage at the gate terminal of M₁ 206, the voltage at the gate terminal of M₁ 206 drops by the same amount as the drop in the voltage at N₅ 212. Because the voltage at the source terminal of M₁ 206 is held at analog ground V_(AG) 118, the gate-to-source voltage of M₁ 206 decreases by the same amount as the drop in the voltage at the gate terminal of M₁ 206. The decrease in the gate-to-source voltage of M₁ 206 causes its drain current to decrease. In response to the decrease in the gate-to-source voltage of M₁ 206 and the decrease in its drain current, the drain-to-source voltage of M₁ 206 increases by a greater magnitude than the decrease in its gate-to-source voltage.

Meanwhile, because the voltage at N₄ 210 is also the voltage at the gate terminal of M₂ 208, the voltage at the gate terminal of M₂ 208 rises by the same amount as the rise in the voltage at N₄ 210. Likewise, because the voltage at N₄ 210 is also the voltage at the gate terminal of M₁₀ 712, the voltage at the gate terminal of M₁₀ 712 rises by the same amount as the rise in the voltage at N₄ 210.

Because the voltage at the source terminal of M₂ 208 is held at analog ground V_(AG) 118, the gate-to-source voltage of M₂ 208 increases by the same amount as the rise in the voltage at the gate terminal of M₂ 208. The increase in the gate-to-source voltage of M₂ 208 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M₂ 208 and the increase in its drain current, the drain-to-source voltage of M₂ 208 decreases by a greater magnitude than the increase in its gate-to-source voltage. Likewise, because the voltage at the source terminal of M₁₀ 712 is held at analog ground V_(AG) 118, the gate-to-source voltage of M₁₀ 712 increases by the same amount as the rise in the voltage at the gate terminal of M₁₀ 712. The increase in the gate-to-source voltage of M₁₀ 712 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M₁₀ 712 and the increase in its drain current, the drain-to-source voltage of M₁₀ 712 decreases by a greater magnitude than the increase in its gate-to-source voltage.

Because the voltage at the source terminal of M₁₀ 712 is held at analog ground V_(AG) 118, the decrease in the drain-to-source voltage of M₁₀ 712 causes the voltage at the voltage at the drain terminal of M₁₀ 712 to drop by the same amount. Because the voltage at the drain terminal of M₁₀ 712 is also the voltage at the gate terminal of M₁₁ 714, the voltage at the gate terminal of M₁₁ 714 drops by the same amount as the drop in the voltage at the drain terminal of M₁₀ 712. Because the voltage at the source terminal of M₁₁ 714 is held at first supply voltage V_(DD) 120, the drop in the voltage at the gate terminal of M₁₁ 714 (i.e., a PMOSFET) causes its gate-to-source voltage to increase by the same amount. The increase in the gate-to-source voltage of M₁₁ 714 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M₁₁ 714 and the increase in its drain current, the drain-to-source voltage of M₁₁ 714 decreases by a greater magnitude than the increase in its gate-to-source voltage.

Because the voltage at the source terminal of M₂ 208 is held at analog ground V_(AG) 118, the voltage at N₅ 212 drops by the same amount as the decrease in drain-to-source voltage of M₂ 208. Thus, the voltage at N₅ 212 drops under the relatively small effect of second current signal i₂ 220 being less than bias current i_(b), and the relatively large effect of the decrease in the drain-to-source voltage of M₂ 208.

Because the voltage at N₄ 210 is also the voltage at the drain terminal of M₁₁ 714 and because the voltage at the source terminal of M₁₁ 714 is held at first supply voltage V_(DD) 120, the voltage at N₄ 210 rises by the same amount as the decrease in the drain-to-source voltage of M₁₁ 714. Furthermore, because the voltage at the source terminal of M₁ 206 is held at analog ground V_(AG) 118, the voltage at N₄ 210 rises by the same amount as the increase in drain-to-source voltage of M₁ 206. Additionally, because the voltage at the source terminal of M₁₁ 714 is held at first supply voltage V_(DD) 120, the voltage at N₄ 210 also rises by the same amount as the decrease in drain-to-source voltage of M₁₁ 714 (i.e., a PMOSFET). Thus, the voltage at N₄ 210 rises under the relatively small effect of first current signal i₁ 218 being greater than bias current i_(b), the relatively large effect of the increase in the drain-to-source voltage of M₁ 206, and the relatively larger effect of the decrease in the drain-to-source voltage of M₁₁ 714.

The increasing of the drain-to-source voltage of M₁ 206 and the decreasing of the drain-to-source voltage of M₂ 208 reinforce each other. The gate-to-source voltage of M₁ 206 decreases with the drain-to-source voltage of M₂ 208 until M₁ 206 is turned OFF.

When M₁ 206 is OFF, it does not conduct current. Without drain current, the decreasing of the gate-to-source voltage of M₁ 206 no longer effects its drain-to-source voltage. However, the voltage at N₄ 210 continues to rise under the relatively small effect of first current signal i₁ 218 being greater than bias current i_(b) and the relatively larger effect of the decrease in the drain-to-source voltage of M₁₁ 714 until the parasitic capacitances at N₄ 210 are fully charged and the voltage at N₄ 210 is HIGH.

It will be recognized that M₁₀ 712 and M₁₁ 714 form a positive feedback loop that amplifies first current signal i₁ 218 and applies an exponentially growing current to the drain terminal of M₁ 206. Thus, the parasitic capacitances at N₄ 210 are charged under the combined effects of first current signal i₁ 218 and the exponentially growing current drawn from first supply voltage V_(DD) 120 by M₁₁ 714.

First vertical latch reset switch 704 and second vertical latch reset switch 708 act to reduce the power dissipated by, respectively, first vertical latch 702 and second vertical latch 706 during the reset phase. For example, when the voltage of Ck.bar 718 is DOWN (i.e., the reset phase), M₁₂ 716 (i.e., a PMOSFET) connects the gate terminal of M₁₁ 714 to first supply voltage V_(DD) 120. With the gate and source terminals of M₁₁ 714 connected together, the gate-to-source voltage of M₁₁ 714 is made to equal zero, holding M₁₁ 714 OFF. This disrupts the latching action of first vertical latch 702 so that cross connected pair 202 can assume a state independent of the state of first vertical latch 702.

However, after the start of the sampling phase, the gate-to-source voltages of M₁₂ 716 and M₁₅ 724 (i.e., PMOSFETs) can drift to values greater than their threshold voltages such that M₁₁ 714 and M₁₄ 722 turn ON. Having M₁ 206, M₂ 208, M₁₀ 712, M₁₁ 714, M₁₃ 720, and M₁₄ 722 all ON before the MOSFETs change states can cause latch circuit 700 to draw a large amount of current. Latch circuit 700 acts, in response to first and second current signals i ₁ 218 and i₂ 220, to force one MOSFET of second cross connected pair 710 (e.g., M₁₆ 726) ON while the other MOSFET of second cross connected pair 710 (e.g., M₁₇ 728) remains OFF. The MOSFET of second cross connected pair 710 (e.g., M₁₆ 726) that turns ON connects the gate terminal of its corresponding vertical latch MOSFET (e.g., M₁₁ 714) to first supply voltage V_(DD) 120. With the gate and source terminals of the corresponding vertical latch MOSFET connected together, the gate-to-source voltage of the corresponding vertical latch MOSFET is made to equal zero, holding the corresponding vertical latch MOSFET OFF. In this manner, second cross connected pair 710 acts to prevent latch circuit 700 from drawing unnecessary current before the MOSFETs change states during the sampling phase.

For example, when, at the start of the sampling phase, first current signal i₁ 218 is slightly larger than bias current i_(b) (i.e., small positive signal current i_(s)), then first current signal i₁ 218 slowly continues to charge the parasitic capacitances at N₄ 210, which causes the voltage at N₄ 210 to rise slightly. Because the voltage at N₄ 210 is also the voltage at the gate terminal of M₁₀ 712, the voltage at the gate terminal of M₁₀ 712 rises by the same amount as the rise in the voltage at N₄ 210.

Because the voltage at the source terminal of M₁₀ 712 is held at analog ground V_(AG) 118, the gate-to-source voltage of M₁₀ 712 increases by the same amount as the rise in the voltage at the gate terminal of M₁₀ 712. The increase in the gate-to-source voltage of M₁₀ 712 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M₁₀ 712 and the increase in its drain current, the drain-to-source voltage of M₁₀ 712 decreases by a greater magnitude than the increase in its gate-to-source voltage. Because the voltage at the source terminal of M₁₀ 712 is held at analog ground V_(AG) 118, the decrease in the drain-to-source voltage of M₁₀ 712 causes the voltage at the drain terminal of M₁₀ 712 to drop by the same amount.

Because the voltage at the drain terminal of M₁₀ 712 is also the voltage at the gate terminal of M₁₆ 726, the voltage at the gate terminal of M₁₆ 726 drops by the same amount as the drop in the voltage at the drain terminal of M₁₀ 712.

Because the voltage at the source terminal of M₁₆ 726 is held at first supply voltage V_(DD) 120, the drop in the voltage at the gate terminal of M₁₆ 726 (i.e., a PMOSFET) causes its gate-to-source voltage to increase by the same amount. The increase in the gate-to-source voltage of M₁₆ 726 causes its drain current to increase. In response to the increase in the gate-to-source voltage of M₁₆ 726 and the increase in its drain current, the drain-to-source voltage of M₁₆ 726 decreases by a greater magnitude than the decrease in its gate-to-source voltage. Because the voltage at the source terminal of M₁₆ 726 is held at first supply voltage V_(DD) 120, the decrease in the drain-to-source voltage of M₁₆ 726 (i.e., a PMOSFET) causes the voltage at the drain terminal of M₁₆ 726 to rise by the same amount.

Because the voltage at the drain terminal of M₁₆ 726 is also the voltage at the gate terminal of M₁₄ 722, the voltage at the gate terminal of M₁₄ 722 rises by the same amount as the rise in the voltage at the drain terminal of M₁₆ 726. Because the voltage at the source terminal of M₁₄ 722 is held at first supply voltage V_(DD) 120, the rise in the voltage at the gate terminal of M₁₄ 722 (i.e. a PMOSFET) causes its gate-to-source voltage to decrease by the same amount.

The decrease in the gate-to-source voltage of M₁₄ 722 ensures that it is less than its threshold voltage so that M₁₄ 722 is held OFF. Having M₁₄ 722 held OFF until first current signal i₁ 218 charges the parasitic capacitances at N₄ 210 to its new steady state voltage of HIGH prevents latch circuit 700 from drawing unnecessary current during the sampling phase.

For an ADC that incorporates an array of comparators based on latch circuit 700, in which the parameters that define latch circuit 700 (i.e., supply voltages, clock frequency, etc.) had specific values, where the ADC was configured with Exclusive OR gates in the manner of the present invention, simulation showed an improvement in the BER from 10⁻⁵⁰ to 10⁻¹⁰⁰. Latch circuit 700 is further described in application Ser. No. 10/083,463, filed on Feb. 27, 2002, which is incorporated herein by reference.

Although the present invention is described in relation to comparators realized with current-mode latch circuits, the skilled artisan will appreciate that the teachings of the present invention are not limited to this embodiment. A signal based on any characteristic (e.g., voltage, resistance, etc.) that indicates that a comparator is in a steady state can be used in an embodiment of the present invention to identify a comparator in a metastable condition. Indeed, such a signal need not be the output of the comparator. Therefore, the present invention is not limited to current-mode latch circuit comparator embodiments.

FIG. 8 shows a flow chart of a method 800 for increasing, in an array of comparators, a rate at which a comparator in a metastable condition transitions to a steady state. In method 800, at a step 802, the comparator in the metastable condition in the array of comparators is identified. At a step 804, a bias current is provided to the identified comparator in the metastable condition, such that the rate at which the comparator in the metastable condition transitions to the steady state is increased. Preferably, the bias current is provided by controlling a current output from a variable current source that provides the bias current for a latch circuit of the identified comparator in the metastable condition.

To further explain step 802, FIG. 9 shows a flow chart of a preferred method to identify the comparator in the metastable condition. At a step 902, a characteristic of a first comparator of the array of comparators is compared with a characteristic of a second comparator of the array of comparators. The first comparator and the second comparator are separated in the array of comparators by a third comparator in the array of comparators. At a step 904, it is determined if the third comparator is the comparator in the metastable condition based on the compared characteristics. Preferably, the characteristics are compared by receiving the characteristics as inputs to an Exclusive OR gate.

In an embodiment, the bias current is provided by controlling a current output from a variable current source that provides the bias current for a latch circuit of the identified comparator in the metastable condition with an output of an Exclusive OR gate.

In another embodiment, the bias current is provided by connecting a first current source in parallel with a second current source to increase the bias current for a latch circuit of the identified comparator in the metastable condition. Preferably, a switch that connects the first current source in parallel with the second current source is controlled by an output of an Exclusive OR gate.

FIG. 10 shows a flow chart of a method 1000 for increasing, in an array of comparators that includes a first, a second, and a third comparator, a rate at which the third comparator transitions to a steady state. In method 1000, at a step 1002, an output of the first comparator is compared with an output of the second comparator. At a step 1004, a bias current is provided to the third comparator.

In an embodiment, the outputs are compared by receiving the first and second outputs as inputs to an Exclusive OR gate. Preferably, a variable current source that provides the bias current for a latch circuit of the third comparator is controlled based on an output of an Exclusive OR gate.

In another embodiment, the bias current is provided to the third comparator by connecting a first current source in parallel with a second current source to increase the bias current for a latch circuit of the third comparator. Preferably, a switch that connects the first current source in parallel with the second current source is controlled based on an output of an Exclusive OR gate.

Preferably, the bias current is provided by controlling a current output from a variable current source that provides the bias current for a latch circuit of the identified comparator in the metastable condition.

CONCLUSION

While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example, and not limitation. It will be apparent to persons skilled in the relevant art that various changes in form and detail can be made therein without departing from the spirit and scope of the invention. Thus the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents. 

1. An array of comparators, comprising: a first, a second, and a third comparator; a first device configured to perform a comparison of a characteristic of said first comparator with said characteristic of said second comparator; and a second device configured to increase, based on said comparison, a rate at which said third comparator transitions from a metastable state to a steady state; wherein said first device is an Exclusive OR gate.
 2. The array of comparators of claim 1, wherein said characteristic is a quantized signal.
 3. An array of comparators, comprising: a first, a second, and a third comparator: a first device configured to perform a comparison of a characteristic of said first comparator with said characteristic of said second comparator; and a second device configured to increase, based on said comparison, a rate at which said third comparator transitions from a metastable state to a steady state; wherein said comparison indicates that said first comparator is in a first steady state and said second comparator is in a second steady state, said second steady state different from said first steady state.
 4. The array of comparators of claim 3, wherein said second device is a current source configured to provide a bias current to said third comparator.
 5. The array of comparators of claim 4, wherein said current source is a variable current source.
 6. The array of comparators of claim 5, wherein said rate at which said third comparator transitions from said metastable state to said steady state is increased by increasing said bias current.
 7. An array of comparators, comprising: a first, a second, and a third comparator; a first device configured to perform a comparison of a characteristic of said first comparator with said characteristic of said second comparator; and a second device configured to increase, based on said comparison, a rate at which said third comparator transitions from a metastable state to a steady state; wherein said second device is a current source configured to provide a bias current to said third comparator and said current source comprises: a first current source; a switch coupled in series with said first current source; and a second current source coupled in parallel with said first current source and said switch.
 8. The array of comparators of claim 7, wherein said rate at which said third comparator transitions from said metastable state to said steady state is increased by increasing said bias current by closing said switch.
 9. In an array of comparators, a system for increasing a rate at which a comparator in a metastable state transitions to a steady state, comprising: a first device configured to determine an identity of the comparator in the metastable state; and a second device configured to increase, based on said identity, the rate at which the comparator in the metastable state transitions to the steady state; wherein said first device is an Exclusive OR gate.
 10. The system of claim 9, wherein said Exclusive OR gate is configured to receive an output from a first comparator in the array of comparators and an output from a second comparator in the array of comparators, said first comparator and said second comparator different from the comparator in the metastable state.
 11. The system of claim 9, wherein said second device is a current source configured to provide a bias current to the comparator in the metastable state.
 12. The system of claim 11, wherein said current source is a variable current source.
 13. The system of claim 12, wherein said rate at which the comparator in the metastable state transitions to the steady state is increased by increasing said bias current.
 14. In an array of comparators, a system for increasing a rate at which a comparator in a metastable state transitions to a steady state, comprising: a first device configured to determine an identity of the comparator in the metastable state; and a second device configured to increase, based on said identity, the rate at which the comparator in the metastable state transitions to the steady state; wherein said second device is a current source configured to provide a bias current to said comparator and said current source comprises: a first current source; a switch coupled in series with said first current source; and a second current source coupled in parallel with said first current source and said switch.
 15. The system of claim 14, wherein said rate at which the comparator in the metastable state transitions to the steady state is increased by increasing said bias current by closing said switch.
 16. An analog to digital converter, comprising: an array of comparators having respective inputs configured to receive an analog signal and respective outputs configured to produce quantized signals responsive to said analog signal; a priority encoder coupled to said array of comparators and configured to produce a digital signal responsive to said quantized signals; an array of first devices coupled to said array of comparators and configured to identify, from said array of comparators, a comparator in a metastable state; and an array of second devices coupled to said array of first devices and configured to increase a rate at which said comparator in said metastable state transitions to a steady state.
 17. The analog to digital converter of claim 16, wherein said first devices are Exclusive OR gates.
 18. The analog to digital converter of claim 16, wherein said second devices are current sources. 